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LT3509 Dual 36V, 700mA Step-Down Regulator DESCRIPTION
The LT(R)3509 is a dual, current mode, step-down switching regulator, with internal power switches each capable of providing 700mA output current. This regulator provides a compact and robust solution for multi-rail systems in harsh environments. It incorporates several protection features including overvoltage lockout and cycle by cycle current limit. Thermal shutdown provides additional protection. The loop compensation components and the boost diodes are integrated on-chip. Switching frequency is set by a single external resistor. External synchronization is also possible. The high maximum switching frequency allows the use of small inductors and ceramic capacitors for low ripple. Constant frequency operation above the AM band avoids interference with radio reception, making the LT3509 well suited for automotive applications. Each regulator has an independent shutdown and soft-start control pin. When both converters are powered down, the common circuitry enters a low current shutdown state.
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.
FEATURES
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Two 700mA Switching Regulators with Internal Power Switches Wide 3.6V to 36V Operating Range Over-Voltage Lockout Protects Circuit Through 60V Supply Transients Short Circuit Robust Low Dropout Voltage - 95% Maximum Duty Cycle Adjustable 300kHz to 2.2MHz Switching Frequency Synchronizable Over the Full Range Uses Small Inductors and Ceramic Capacitors Integrated Boost Diodes Internal Compensation Thermally Enhanced 14 Lead (4 mm x 3 mm) DFN and 16 Lead MSOP Packages
APPLICATIONS
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Automotive Electronics Industrial Controls Wall Transformer Regulation Networking Devices CPU, DSP, or FPGA Power
TYPICAL APPLICATION
3.3V and 5V Dual Output Step-Down Converter
6.5V TO 36V (Transient to 60V) 90 2.2F VIN BOOST1 3.3V 700mA 6.8H 0.1F SW1 LT3509 31.6k MBRM140 10F 10.2k 1nF DA1 FB1 DA2 FB2 MBRM140 53.6k 22F 60 1nF 60.4k fSW = 700kHz 10.2k
3509 TA01a 3509 TA01b
Efficiency
BD BOOST2 SW2 5V 700mA EFFICIENCY (%) 0.1F 10H
85 80
VOUT = 5V VOUT = 3.3V
75 70 65
RUN/SS1 RUN/SS2 SYNC GND RT
55 50 0.0
VIN = 12V fSW = -700kHz 0.1 0.2 0.3 0.4 0.5 LOAD CURRENT (A) 0.6 0.7
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(Note 1)
ABSOLUTE MAXIMUM RATINGS
VIN Pin (Note 2) ........................................................60V BD Pin .......................................................................20V BOOST Pins ..............................................................60V BOOST Pins above SW .............................................30V RUN/SS, FB, RT, SYNC pins ........................................6V Operating Junction Temperature Range (Notes 3, 5) LT3509E ............................................. -40C to 125C LT3509I .............................................. -40C to 125C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec.) MSE16 Package ................................................ 300C
PIN CONFIGURATION
TOP VIEW DA1 BOOST1 SW1 VIN SW2 BOOST2 DA2 1 2 3 4 5 6 7 15 14 FB1 13 RUN/SS1 12 BD 11 SYNC 10 RT 9 RUN/SS2 8 FB2 DA1 BOOST1 SW1 VIN VIN SW2 BOOST2 DA2 1 2 3 4 5 6 7 8 TOP VIEW 16 15 14 13 12 11 10 9 FB1 RUN/SS1 AGND BD SYNC RT RUN/SS2 FB2
17
DE14 PACKAGE 14-LEAD (4mm 3mm) PLASTIC DFN JA = 43C/W, JC = 4.3C/W EXPOSED PAD (PIN 15) IS GND, MUST BE SOLDERED TO PCB
MSE PACKAGE 16-LEAD PLASTIC MSOP JA = 43C/W, JC = 4.3C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LT3509EDE#PBF LT3509IDE#PBF LT3509EMSE#PBF LT3509IMSE#PBF TAPE AND REEL LT3509EDE#TRPBF LT3509IDE#TRPBF LT3509EMSE#TRPBF LT3509IMSE#TRPBF PART MARKING* 3509 3509 3509 3509 PACKAGE DESCRIPTION 14-Lead (4mm x 3mm) Plastic DFN 14-Lead (4mm x 3mm) Plastic DFN 16-Lead Plastic MSOP with Exposed Pad 16-Lead Plastic MSOP with Exposed Pad TEMPERATURE RANGE -40C to 125C -40C to 125C -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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ELECTRICAL CHARACTERISTICS
PARAMETER VIN Undervoltage Lockout VIN Overvoltage Lockout Input Quiescent Current Input Shutdown Current Feedback Pin Voltage Reference Voltage Line Regulation RUN/SS Shutdown Threshold RUN/SS Voltage for Full IOUT RUN/SS Pin Pull-up Current Feedback Pin Bias Current Switch Current Limit DA Comparator Current Threshold Boost Pin Current Switch Leakage Current Switch Saturation Voltage Minumum Boost Voltage above Switch Boost Diode Forward Voltage Boost Diode Leakage Switching Frequency ISW = 0.9A
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C, VIN = 12V, VBD = 5V. (Note 3)
CONDITIONS MIN 37 Not Switching VFB > 0.8V V(RUN/SS[1,2]) < 0.3V
l
TYP 3.3 38.5 1.9 9
MAX 3.6 40 2.2 15 0.816 0.8 2 1.3 500 1.9 1.2 36 1.0 2.2 0.9 5 1.08 290 2.5 150
UNITS V V mA A V %/V V V A nA A A mA A V V V A MHz kHz MHz ns
0.784 0.4 0.7
0.8 0.01 0.6 1 90 1.4 0.95 22 0.01 0.32 1.5 0.7 0.1
3.6V < VIN < 36V
l l
1.05 0.7
ISW = 0.9A (Note 4) ISW = 0.9A IBD= 20mA VR = 30V RT = 40.2k RT = 180k RT = 14.1k
l
0.92 237 2.0
1.0 264 2.2 80
Switch Minimum Off Time Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2. Absolute Maximum Voltage at the VIN pin is 60V for non-repetitive 1 second transients and 36V for continuous operation. Note 3. The LT3509E is guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization
l
and correlation with statistical process controls. The LT3509I is guaranteed over the full -40C to 125C temperature range. Note 4. Switch Saturation Voltage is guaranteed by design. Note 5. This IC includes over-temperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating temperature when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current VOUT = 5V, fSW = 2.0MHz
95 90 VIN = 12V 85 EFFICIENCY (%) EFFICIENCY (%) 80 VIN = 24V 75 70 65 60 TA = 25C 95 90 85 80 75 70 65 60 55 0 0.2 0.4 ILOAD(A)
3509 G01
Efficiency vs Load Current VOUT = 3.3V, fSW = 2.0MHz
TA = 25C 90 85 VIN = 12V 80 EFFICIENCY (%) 75 70 65 60 55 0 0.2 0.4 ILOAD(A)
3509 G02
Efficiency vs Load Current VOUT = 1.8V, fSW = 0.7MHz
TA = 25C
VIN = 12V
0.6
0.8
0.6
0.8
50
0
0.2
0.4 ILOAD(A)
0.6
0.8
3509 G03
Switch VCE(SAT) vs ISW
0.35 0.3 20 0.25 IBOOST (mA) VCE(SAT) (V) 0.2 0.15 0.1 5 0.05 0 0 15 TA = 25C 25
IBOOST vs ISW
TA = 25C
10
0
0.2
0.4
0.6 ISW(A)
0.8
1.0
3509 G04
0
0.2
0.4
0.6 ISW(A)
0.8
1
3509 G05
Boost Diode Characteristics
1.2 1 FREQUENCY (MHz) 0.8 Vf (V) 0.6 0.4 0.2 0 0 1.5 TA = 25C 2.2 2.0
Frequency vs RT
TA = 25C
1.0
0.5
0
100 BOOST DIODE CURRENT (mA)
50
150
3509 G06
0
20
40
60
80 100 120 140 160 180 RT(k)
3509 G07
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TYPICAL PERFORMANCE CHARACTERISTICS
FSW vs Temperature
1.025 1.02 1.015 1.01 FSW (MHz) FBREF (V) 1.005 1 0.995 0.99 0.985 0.98 0.975 -50 -25 0 75 TEMPERATURE(C) 25 50 100 125 0.79 -50 -25 0 0 75 TEMPERATURE(C) 25 50 100 125 0 0.2 0.4 ILOAD (A) 0.6 0.8
3509 G10
FB Pin Voltage vs. Temperature
0.81 30 25 0.805 MAXIMUM VIN (V) 20 15 10 5
Max VIN for Constant Frequency VOUT = 3.3V, fSW = 2MHz
TA = 25C TA = 85C
RT = 40.2k
0.8
0.795
3509 G08
3509 G09
Max VIN for Constant Frequency VOUT = 5V, fSW = 2MHz
45 40 35 MAXIMUM VIN (V) 30 25 20 15 10 5 0 0 0.2 0.4 ILOAD (A)
3509 G11
Min ON Time vs. Temperature ILOAD = 0.3A
140 120 MINIMUM ON TIME (ns) 100 80 60 40 20 0 -50 -25
TA = 25C TA = 85C
0.6
0.8
0
25 50 75 TEMPERATURE(C)
100
125
3509 G12
ILIM vs Temperature
2 SWITCH 1.5 ILIM (A) ILIM (A) 1.4 1.2 1 0.8 0.6 0.5 0.4 0.2 0 -40 0 0 80 TEMPERATURE (C) 40 120
3509 G13
ILIM vs Duty Cycle
1.8 1.6 TA = 25C
1 DA
0
10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
3509 G14
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PIN FUNCTIONS
(DFN/MSE)
DA1, DA2 (Pins 1, 7 / Pins 1, 8): The DA pins are the anode connections for the catch diodes. These are connected internally to the exposed ground pad by current sensing resistors. BOOST1, BOOST2 (Pins 2, 6 / Pins 2, 7): The BOOST pins are used to dynamically boost the power transistor base above VIN to minimize the voltage drop and power loss in the switch. These should be tied to the associated switch pins through the boost capacitors. SW1, SW2 (Pins 3, 5 / Pins 3, 6): The SW pins are the internal power switch outputs. These should be connected to the associated inductors, catch diode cathodes, and the boost capacitors. VIN (Pin 4 / Pins 4, 5): The VIN pins supply power to the internal power switches and control circuitry. In the MSE package the VIN pins must be tied together. The input capacitor should be placed as close as possible to the supply pins. FB1, FB2 (Pins 14, 8 / Pins 16, 9): The FB pins are used to set the regulated output voltage relative to the internal reference. These pins should be connected to a resistor divider from the regulated output such that the FB pin is at 0.8V when the output is at the desired voltage. RUN/SS1, RUN/SS2 (Pins 13, 9 / Pins 15, 10): The RUN/SS pins enable the associated regulator channel. If both pins are pulled to ground, the device will shut-down to a low power state. In the range 0.7V to 2.0V, the regulators are enabled but the peak switch current and the DA pin maximum current are limited to provide a soft-start function. Above 2V, the full output current is available. The inputs incorporate a 1A pull-up so that they will float high or charge an external capacitor to provide a current limited
soft-start. The pins are pulled down by approximately 250A in the case of overvoltage or overtemperature conditions in order to discharge the soft-start capacitors. The pins can also be driven by a logic control signal of up to 5.0V. In this case, it is necessary place a 10k to 50k resistor in series along with a capacitor from the RUN/SS pin to ground to ensure that there will be a soft-start for both initial turn on and in the case of fault conditions. Do not tie these pins to VIN. RT (Pin 10 / Pin 11): The RT pin is used to set the internal oscillator frequency. A 40.2k resistor from RT to ground results in a nominal frequncy of 1MHz. SYNC (Pin 11 / Pin12): The SYNC pin allows the switching frequency to be synchronized to a external clock. Choose RT resistor to set a free-run frequency at least 12% less than the external clock frequency for correct operation. BD (Pin 12 / Pin 13): The BD pin is common anode connection of the internal Schottky boost diodes. This provides the power for charging the BOOST capacitors. It should be locally bypassed for best performance. GND (Exposed Pad): This is the reference and supply ground for the regulator. The exposed pad must be soldered to the PCB and electrically connected to supply ground. Use a large ground plane and thermal vias to optimize thermal performance. The current in the catch diodes also flows through the GND pad to the DA pins. AGND (Pin 14, MSE Package Only): This is the connected to the ground connection of the chip and may be used as a separate return for the low current control side components. It should not be used as the only ground connection or as a connection return for load side components.
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BLOCK DIAGRAM
COMMON CIRCUITRY VIN BD C1 OVERVOLTAGE DETECT BOOST DIODE BOOST SHUTDOWN AND SOFT-START CONTROL C3 C2 RUN/SS2 MAIN CURRENT COMPARATOR POWER SWITCH SWITCH LOGIC SWITCH DRIVER SW VREF AND CORE VOLTAGE REGULATOR SYNC RT RT OSCILLATOR DA CURRENT COMPARATOR -17mV VC DA ERROR AMPLIFIER R1 VREF 0.8V 18m FB R2 GND
3509 BD
1 OF 2 REGULATOR CHANNELS SHOWN
NOTE: THE BD PIN IS COMMON TO BOTH CHANNELS.
RUN/SS1
C4 L1 VOUT D1 C5
-
SLOPE
CLAMP
Figure 1. Functional Block Diagram
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Overview
OPERATION
The LT3509 is a dual, constant frequency, current mode switching regulator with internal power switches. The two independent channels share a common voltage reference and oscillator and operate in phase. The switching frequency is set by a single resistor and can also be synchronized to an external clock. Operation can be best understood by referring to the Block Diagram (Figure 1). Startup and Shutdown When the RUN/SS[1,2] pins are pulled low (<0.3V) the associated regulator channel is shut-down. If both channels are shut down, the common circuitry also enters a low current state. When the RUN/SS pins exceed approximately 0.7V, the common circuitry and the associated regulator are enabled but the output current is limited. From 0.7V up to 2.0V the current limit increases until it reaches the full value. The RUN/SS pins also incorporate a 1A pull-up to approximately 3V, so the regulator will run if they are left open. A capacitor to ground will cause a current limited soft-start to occur at power-up. In the case of undervoltage, overvoltage or over-temperature conditions the internal circuitry will pull the RUN/SS pins down with a current of approximately 250A. Thus a new soft-start cycle will occur when the fault condition ends. Voltage and Current Regulation The power switches are controlled by a current-mode regulator architecture. The power switch is turned on at the beginning of each clock cycle and turned off by the Main Current Comparator. The inductor current will ramp up while the switch is on until it reaches the peak current threshold. The current at which it turns off is determined by the Error Amp and the internal compensation network. When the switch turns off, the current in the inductor will cause the SW pin to fall rapidly until the catch diode, D1, conducts. The voltage applied to the inductor will now reverse and the current will linearly fall. The resistor divider, R1 and R2, sets the desired output voltage such that when the voltage at FB reaches 0.8V, the Main Current Comparator threshold will fall and reduce the peak inductor current and hence the average current, until it matches the load current. By making current the controlled variable in the loop, the inductor impedance is effectively removed from the transfer function and the compensation network is simplified. The Main Current Comparator threshold is reduced by the slope compensation signal to eliminate sub-harmonic oscillations at duty cycles >50%. Current Limiting Current mode control provides cycle by cycle current limiting by means of a clamp on the maximum current that can be provided by the switch. A comparator monitors the current flowing through the catch diode via the DA pin. This comparator delays switching if the diode current is higher than 0.95A (typical). This current level is indicative of a fault condition such as a shorted output with a high input voltage. Switching will only resume once the diode current has fallen below the 0.95A limit. This way the DA comparator regulates the valley current of the inductor to 0.95A during a short circuit. This will ensure the part will survive a short circuit event. Over and Under Voltage Shutdown A basic under voltage lockout prevents switching if VIN is below 3.3V (typical). The overvoltage shutdown stops the part from switching when VIN is greater than 38.5V (typical). This protects the device and its load during momentary overvoltage events. After the input voltage falls below 38.5V, the part initiates a soft start sequence and resumes switching. BOOST Circuit To ensure best efficiency and minimum dropout voltage the output transistor base drive is boosted above VIN by the external boost capacitors (C4). When the SW pin is low the capacitors are charged via the BOOST diodes and the supply on BD.
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Setting The Output Voltage The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the resistors according to: V R1= R2 * OUT - 1 0.8 The designators correspond to Figure 1. R2 should be 20k or less to avoid bias current errors. Frequency Setting The timing resistor RT for any desired frequency in the range 270kHz to 2.2MHz can be calculated from the following formula: 1.166 RT = - 0.166 * 40.2 fSW Where fSW is in MHz and RT is in k.
Table 1. Standard E96 Resistors for Common Frequencies
FREQUENCY TIMING RESISTOR RT (k) 165 150 113 88.7 40.2 16.2 14 270 kHz 300 kHz 400kHz 500kHz
APPLICATIONS INFORMATION
Shutdown and Soft Start When the RUN/SS pins are pulled to ground, the part will shut down to its lowest current state of approximately 10A. If driving a large capacitive load it may be desirable to use the current limiting soft start feature. Connecting capacitors to ground from the RUN/SS pins will control the delay until full current is available. The pull-up current is 1A and the full current threshold is 2V so the start-up time is given by: T = 2 x C x 106 s For example a 0.005F capacitor will give a time to full current of 10ms. If both outputs can come up together then the two inputs can be paralleled and tied to one capacitor. In this case use twice the capacitor value to obtain the same start-up time. During the soft-start time both the peak current threshold and the DA current threshold will track so the part will skip pulses as required to limit the maximum inductor current. Starting up into a large capacitor is not much different to starting into a shortcircuit in this respect.
VSW 10V/DIV
IL 0.2A/DIV VOUT 5V/DIV TIME 1ms/DIV
3509 F02
1MHz 2MHz 2.2MHz
Figure 2. Soft-Start
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APPLICATIONS INFORMATION
External Synchronization The external synchronization provides a trigger to the internal oscillator. As such, it can only raise the frequency above the free-run value. To allow for device and component tolerances, the free run frequency should be set to at least 12% lower than the lowest supplied external synchronization reference. The oscillator and hence the switching frequency can then pushed up from 12% above the free-run frequency, set by the selected RT. For example, if the minimum external clock is 300kHz, the RT should be chosen for 264KHz. The SYNC input has a threshold of 1.0V nominal so it is compatible with most logic levels. The duty cycle is not critical provided the high or low pulse width is at least 80ns. Design Procedure Before starting detailed design a number of key design parameters should be established as these may affect design decisions and component choices along the way. One of the main things to determine apart from the desired output voltages is the input voltage range. Both the normal operating range and the extreme conditions of surges and/or dips or brown-outs need to be known. Then the operating frequency should be considered and if there are particular requirements to avoid interference. If there are very specific frequencies that need to be avoided then external synchronization may be needed. This could also be desirable if multiple switchers are used as low frequency beating between similar devices can be undesirable. For efficient operation this converter requires a boost supply so that the base of the output transistor can be pumped above the input voltage during the switch on time. Depending on the input and output voltages the boost supply can be provided by the input voltage, one of the regulated outputs or an independent supply such as an LDO. Input Voltage Range Firstly, the LT3509 imposes some hard limits due to the undervoltage lock-out and the overvoltage protection. A given application will also have a reduced, normal operating range over which maximum efficiency and lowest ripple are obtained. This usually requires that the device is operating at a fixed frequency without skipping pulses. There may also be zones above and below the normal range where regulation is maintained but efficiency and ripple may be compromised. At the low end, insufficient input voltage will cause loss of regulation and increased ripple-this is the dropout range. At the high end if the duty cycle becomes too low this will cause pulse skipping and excessive ripple. This is the pulse-skip region. Both situations also lead to higher noise at frequencies other than the chosen switching frequency. Occasional excursions into pulse-skip mode, during surges for example, may be tolerable. Pulse skipping will also occur at light loads even within the normal operating range but ripple is usually not degraded because at light load the output capacitor can hold the voltage steady between pulses. To ensure the regulator is operating in continuous mode it is necessary to calculate the duty cycle for the required output voltage over the full input voltage range. This must then be compared with minimum and maximum practical duty cycles.
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The minimum on time increases with increasing temperature so the value for the maximum operating temperature should be used. See the Minimum ON Time vs Load graph in the Typical Performance Characteristics. The maximum input voltage for this duty cycle is given by: VIN(MAX ) = VOUT + VF - VF + VSW DCMIN
APPLICATIONS INFORMATION
In any step-down switcher the duty cycle when operating in continuous, or fixed frequency, mode is dependent on the step-down ratio. This is because for a constant average load current the decay of the inductor current when the switch is off must match the increase in inductor current when the switch is on. The can be estimated by the following formula: DC = VOUT + VF VIN - VSW + VF
Where: DC = Duty Cycle (Fraction of Cycle when Switch is On) VOUT = Output Voltage VIN = Input Voltage VF = Catch Diode Forward Voltage VSW = Switch Voltage Drop Note: This formula neglects switching and inductor losses so in practice the duty cycle may be slightly higher. It is clear from this equation that the duty cycle will approach 100% as the input voltage is reduced and become smaller as the input voltage increases. There are practical limits to the minimum and maximum duty cycles for continuous operation due to the switch minimum off and on times. These are independent of operating frequency so it is clear that range of usable duty cycle is inverserly proportional to frequency. Therefore at higher frequency the input voltage range (for constant frequncy operation) will narrow. The minimum duty cycle is given by: DCMIN = fSW * t ON(MIN) Where: fSW = Switching Frequency tMINON = Switch Minimum on Time
Above this voltage the only way the LT3509 can maintain regulation is to skip cycles so the effective freqeuncy will reduce. This will cause an increase in ripple and the switching noise will shift to a lower frequency. This calculation will in practice drive the maximum switching frequency for a desired step-down ratio.
VOUT 100mV/DIV (AC COUPLED) IL 0.5A/DIV
TIME 1s/DIV
3509 F03
Figure 3. Continuous Mode
VOUT 100mV/DIV (AC COUPLED) IL 0.5A/DIV
TIME 1s/DIV
3509 F04
Figure 4. Pulse Skipping
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APPLICATIONS INFORMATION
Minimum Input Voltage and Boost Architecture The minimum operating voltage is determined either by the LT3509's internal undervoltage lockout of ~3.6V or by its maximum duty cycle. The maximum duty cycle for fixed frequency operation is given by: DCMAX = 1- tOFF(MIN) * fSW It follows that: VIN(MIN) = VOUT + VF - VF + VSW DCMAX boost capacitors to fully saturate the switch. This is most problematic when the BD pin is supplied from the regulated output. The net result is that a higher input voltage will be required to start up the boost system. The typical minimum input voltage over a range of loads is shown in Figure 5 for 3.3V and Figure 6 for 5.0V. When operating at such high duty cycles the peak currents in the boost diodes are greater and this will require a the BD supply to be somewhat higher than would be required at less extreme duty cycles. If operation at low input/output ratios and low BD supply voltages is required it may be desirable to augment the internal boost diodes with external discrete diodes in parallel. Boost Pin Considerations The boost capacitor, in conjunction with the internal boost diode, provides a bootstrapped supply for the power switch that is above the input voltage. For operation at 1MHz and above and at reasonable duty cycles a 0.1F capacitor will work well. For operation at lower frequencies and/or higher duty cycles something larger may be needed. A good rule of thumb is: VIN(MIN) = VOUT + VF - VF + VSW DCMAX
If a reduction in switching frequency can be tolerated the minimum input voltage can drop to just above output voltage. Not only is the output transistor base pumped above the input voltage by the boost capacitor, the switch can remain on through multiple switching cycles resulting in a high effective duty cycle. Thus, this is a true low-dropout regulator. As it is necessary to recharge the boost capacitor from time to time, a minimum width off-cycle will be forced occasionally to maintain the charge. Depending on the operating frequency, the duty cycle can reach 97-98%, although at this point the output pulses will be at a sub-multiple of the programmed frequency. One other consideration is that at very light loads or no load the part will go into pulse skipping mode. The part will then have trouble getting enough voltage on to the
5.5 5 4.5 VIN TO START (V) 4 3.5 3 2.5 2 0.001 TO RUN TO START
where fSW is in MHz and CBOOST is in F
7 TO START 6.5 VIN TO START (V) 6 TO RUN 5.5 5 4.5 4 0.001
0.1 0.01 LOAD CURRENT (A)
1
3509 F05
0.1 0.01 LOAD CURRENT (A)
1
3509 F06
Figure 5. Minimum VIN for 3.3V VOUT
Figure 6. Minimum VIN for 5.0V VOUT
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LT3509
In any case, be sure that the maximum voltage at the BOOST pin is less than 60V and the voltage difference between the BOOST and SW pins is less than 30V. Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L = ( VOUT + VF ) * 2.1MHz fSW
APPLICATIONS INFORMATION
Boost Pin Considerations Figure 7 through Figure 9 show several ways to arrange the boost circuit. The BOOST pin must be more than 2.0V above the SW pin for full efficiency. For outputs of 3.3V and higher, the standard circuit Figure 7 is best. For lower output voltages, the boost diode can be tied to the input Figure 8. The circuit in Figure 7 is more efficient because the boost pin current comes from a lower voltage source. Finally, as shown in Figure 9, the BD pin can be tied to another source that is at least 3V. For example, if you are generating 3.3V and 1.8V, and the 3.3V is on whenever the 1.8V is on, the 1.8V boost diode can be connected to the 3.3V output.
BD VIN VIN BOOST CBOOST LT3509 SW CIN D1 GND DA COUT L1 VOUT
where VF is the voltage drop of the catch diode (~0.5V) and L is in H. The inductor's RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.15. Table 2 lists several vendors and types that are suitable. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3509 limits its switch current in order to protect itself and the system from over-current faults. Therefore, the maximum output current that the LT3509 will deliver depends on the switch current limit, the inductor value and the input and output voltages.
3509 F07
VBOOST - VSW VOUT MAX VBOOST VIN VOUT VOUT 3V
Figure 7. BD Tied to Regulated Output
VBD BOOST LT3509 SW CIN D1 GND DA COUT VIN CBOOST L1 VOUT CIN D1 GND DA COUT BD VIN BOOST CBOOST LT3509 SW L1 VOUT
BD VIN VIN
3509 F09 3509 F08
VBOOST - VSW VIN MAX VBOOST 2VIN
VBOOST - VSW VBD MAX VBOOST VIN VBD VBD 3V
Figure 8. Supplied from VIN
Figure 9. Separate Boost Supply
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APPLICATIONS INFORMATION
When the switch is off, the potential across the inductor is the output voltage plus the catch diode forward voltage. This gives the peak-to-peak ripple current in the inductor: IL = (1- DC) where: DC = Duty Cycle fSW = switching frequency L = inductor value VF = diode forward voltage. The peak inductor and switch current is: ISWPK = ILPK = IOUT + IL 2 VOUT + VF L * fSW There is a limit to the actual minimum duty cycle imposed by the minimum on time of the switch. For a robust design it is important that inductor that will not saturate when the switch is at its minimum on time, the input voltage is at maximum and the output is short-circuited. In this case the full input voltage, less the drop in the switch, will appear across the inductor. This doesn't require an actual short, just starting into a capacitive load will provide the same conditions. The Diode current sensing scheme will ensure that the switch will not turn-on if the inductor current is above the DA current limit threshold, which has a maximum of 1.1A. The peak current under short-circuit conditions can then be calculated from: IPEAK = VIN * tON(MIN) L + 1.1A
To maintain output regulation, this peak current must be less than the LT3509's switch current limit ILIM. This is dependent on duty cycle due to the slope compensation. For ILIM is at least 1.4A at low duty cycles and decreases linearly to 1.0A at DC = 0.8. The theoretical minimum inductance can now be calculated as: LMIN = 1- DCMIN VOUT + VF * f ILIM - IOUT
The inductor should have a saturation current greater than this value. For safe operation with high input voltages this can often mean using a physically larger inductor as higher value inductors often have lower saturation currents for a given core size. As a general rule the saturation current should be at least 1.8A to be short-circuit proof. However, it's generally better to use an inductor larger than the minimum value. The minimum inductor has large ripple currents which increase core losses and require large output capacitors to keep output voltage ripple low. Select an inductor greater than LMIN that keeps the ripple current below 30% of ILIM.
Where DCMIN is the minimum duty cycle called for by the application i.e. DCMIN = VOUT(MAX ) + VF VIN(MIN) - VSW + VF
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The prior analysis is valid for continuous mode operation (IOUT > ILIM / 2). For details of maximum output current in discontinuous mode operation, see Linear Technology's Application Note AN44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid subharmonic oscillations. This minimum inductance is L MIN = ( VOUT + VF )* 1.4 fSW
APPLICATIONS INFORMATION
Table 2. Recommended Inductors
MANUFACTURER/ PART NUMBER Coilcraft LPS4018-222ML LPS5030-332ML LPS5030-472ML LPS6225-682ML LPS6225-103ML Sumida CDRH4D22/HP-2R2N CDRH4D22/HP-3R5N CDRH4D22/HP-4R7N CDRH5D28/HP-6R8N CDRH5D28/HP-8R2N CDRH5D28R/HP-100N Cooper SD52-2R2-R SD52-3R5-R SD52-4R7-R SD6030-5R8-R SD7030-8R0-R SD7030-100-R Toko A997AS-2R2N A997AS-3R3N A997AS-4R7M Wurth 7447745022 7447745033 7447745047 7447745076 7447445100 2.2 3.3 4.7 7.6 10 3.5 3.0 2.4 1.8 1.6 0.036 0.045 0.057 0.095 0.12 2.0 2.0 2.0 2.0 2.0 2.2 3.3 4.7 1.6 1.2 1.07 0.06 0.07 0.1 1.8 1.8 1.8 2.2 3.5 4.7 5.8 8.0 10.0 2.30 1.82 1.64 1.8 1.85 1.7 0.0385 0.0503 0.0568 0.045 0.058 0.065 2.0 2.0 2.0 3.0 3.0 3.0 2.2 3.5 4.7 6.8 8.2 10 3.2 2.5 2.2 3.1 2.7 2.45 0.0035 0.052 0.066 0.049 0.071 0.074 2.4 2.4 2.4 3.0 3.0 3.0 2.2 3.3 4.7 6.8 10 2.8 2.5 2.5 2.7 2.1 0.07 0.066 0.083 0.095 0.105 1.7 2.9 2.9 2.4 2.4 VALUE (H) ISAT (A) DCR () HEIGHT (mm)
where fSW is in MHz and LMIN is in H. If using external synchronization, calculate LMIN using the RT frequency and not the SYNC frequency. Frequency Compensation The LT3509 uses current mode control to regulate the output, which simplifies loop compensation and allows the necessary filter components to be integrated. The fixed internal compensation network has been chosen to give stable operation over a wide range of operating conditions but assumes a minimum load capacitance. The LT3509 does not depend on the ESR of the output capacitor for stability so the designer is free to use ceramic capacitors to achieve low output ripple and small PCB footprint. Figure 10 shows an equivalent circuit for the LT3509 control loop. The error amp is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor is modeled as a transconductance amplifier generating an output current proportional to the voltage at the COMP-NODE. The gain of the power stage (gmp) is 1.1S. Note that the output capacitor integrates this current and that the internal capacitor integrates the error amplifier output current, resulting in two poles in the loop. In most cases, a zero is required and comes either from the output capacitor ESR
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APPLICATIONS INFORMATION
or from RC. This model works well as long as the inductor current ripple is not too low (IRIPPLE > 5% IOUT) and the loop crossover frequency is less than fSW/5. An optional phase lead capacitor (CPL) across the feedback divider may improve the transient response.
1.1S VIN CPL R1 COMPNODE RC 75k VREF = 0.8V 95pF 1.73M R2 COUT 260S LT3509 VOUT
For ceramic capacitors where low capacitance value is more significant than ESR: VRIPPLE = IL / (8 * fSW * COUT ) For electrolytic capacitors where ESR is high relative to capacitive reactance: VRIPPLE = IL * ESR where IL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = IL / 12 Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor transfers to the output, the resulting voltage step should be small compared to the regulation voltage. For a 5% overshoot, this requirement indicates: COUT > 10 * L *(ILIM / VOUT)2 The low ESR and small size of ceramic capacitors make them the preferred type for LT3509 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types.
Figure 10. Small Signal Equivalent Circuit
Output Capacitor Selection The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilize the LT3509's control loop. Because the LT3509 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equations:
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to give insight to an empirical design. Figure 11 shows some load step responses with differing output capacitors and CPL combinations. Input Capacitor The input capacitor needs to supply the pulses of charge demanded during the on time of the switches. Little total capacitance is required as a few hundred millivolts of ripple at the VIN pin will not cause any problems to the device. When operating at 2MHz and 12V, 2F will work well. At the lowest operating frequency and/or at low input voltages a larger capacitor such as 4.7F is preferred.
ILOAD 700mA 300mA
APPLICATIONS INFORMATION
The value of the output capacitor greatly affects the transient response to a load step. It has to supply extra current demand or absorb excess current delivery until the feedback loop can respond. The loop response is dependent on the error amplifier transconductance, the internal compensation capacitor and the feedback network. Higher output voltages necessarily require a larger feedback divider ratio. This will also reduce the loop gain and slow the response time. Fortunately this effect can be mitigated by use of a feed-forward capacitor CPL across the top feedback resistor. The small signal model shown in Figure 10 can be used to model this in a simulator or
ILOAD 700mA 300mA
VOUT (AC) 50mV/DIV
VOUT (AC) 50mV/DIV
TIME 20s/DIV COUT = 10F CPL = 0
TIME 20s/DIV COUT = 10F CPL = 82pF
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Figure 11. Transient Load Response with Different Combinations of COUT and CPL Load Current Step from 300mA to 700mA R1 = 10k, R2 = 32.4k, VIN = 12V, VOUT = 3.3V, fSW = 2.0MHz
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APPLICATIONS INFORMATION
Diode Selection The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID( AVG) = IOUT ( VIN - VOUT ) / VIN The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current limit. If transient input voltages exceed 40V, use a Schottky diode with a reverse voltage rating of 45V or higher. If the maximum transient input voltage is under 40V, use a Schottky diode with a reverse voltage rating greater than the maximum input voltage. Table 3 lists several Schottky diodes and their manufacturers:
Table 3. Schottky Diodes
MANUFACTURER/ PART NUMBER On Semiconductor MBRM140 MicroSemi UPS140 Diodes Inc. DFLS140L 1N5819HW 40 40 1 1 550 450 40 1 450
D2 VIN CIN RUN/SS1 VIN BD BOOST LT3509 SW D1 RUN/SS2 DA GND COUT CBOOST L1 VOUT
VR (V) 40
IAVE (A) 1
VF at 1A (mV) 550
providing the boost supply to the BD pin. In this case the voltage drop of the other switch will increase and lower the efficiency. This could eventually cause the part to reach the thermal shutdown limit. One other important feature of the part that needs to be considered is that there is a parasitic diode in parallel with the power switch. In normal operation this is reverse biased but it could conduct if the load can be powered from an alternate source when the LT3509 has no input. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with one of the LT3509 regulated outputs. If the SW pin is at more than about 4V the VIN pin can attain sufficient voltage for LT3509 control circuitry to power-up to the quiescent bias level and up to 2mA could be drawn from the backup supply. This can be minimized if some discrete FETs or open drain buffers are used to pull down the RUN/SS pins. Of course the gates need to be driven from the standby or battery backed supply. If there is the possibility of a short-circuit at the input or just other parallel circuits connected to VIN it would be best to add a protection diode in series with VIN. This will also protect against a reversed input polarity. These concepts are illustrated in Figure 12.
Short and Reverse Protection Provided the inductors are chosen to not go deep into their saturation region at the maximum ILIMIT current the LT3509 will tolerate a short-circuit on one or both outputs. The excess current in the inductor will be detected by the DA comparator and the frequency will reduced until the valley current is below the limit. This shouldn't affect the other channel unless the channel that is shorted is also
SLEEP
3509 F12
Figure 12. Reverse Bias Protection
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* The loop from the regulated outputs through the output capacitor back to the ground plane. Excess impedance here will result in excessive ripple at the output. The area of the SW and BOOST nodes should as small as possible. Also the feedback components should be placed as close as possible to the FB pins so that the traces are short and shielded from the SW and BOOST nodes by the ground planes. Figure 13 shows a detail view of a practical board layout showing just the top layer. The complete board is somewhat larger at 7.5cm x 7.5cm. The device has been evaluated on this board in still air running at 700kHz switching frequency. One channel was set to 5V and the other to 3.3V and both channels were fully loaded to 700mA. The device temperature reached approximately 15C above ambient for input voltages below 12V. At 24V input it was slightly higher at 17 above ambient.
APPLICATIONS INFORMATION
Hot Plugging Considerations The small size, reliability and low impedance of ceramic capacitors make them attractive for the input capacitor. Unfortunately they can be hazardous to semiconductor devices if combined with an inductive supply loop and a fast power transition such as through a mechanical switch or connector. The low-loss ceramic capacitor combined with the just a small amount of wiring inductance forms an underdamped resonant tank circuit and the voltage at the VIN pin of the LT3509 can ring to twice the nominal input voltage. See Linear Technology Application Note 88 for more details. PCB Layout and Thermal Design The PCB layout is critical to both the electrical and thermal performance of the LT3509. Most important is the connection to the exposed pad which provides the main ground connection and also a thermal path for cooling the chip. This must be soldered to a topside copper plane which is also tied to backside and/or internal plane(s) with an array of thermal vias. To obtain the best electrical performance particular attention should be paid to keeping the following current paths short: * The loop from the VIN pin through the input capacitor back to the ground pad and plane. This sees high di/dt transitions as the power switches turn on an off. Excess impedance will degrade the minimum usable input voltage and could cause crosstalk between channels. * The loops from the switch pins to the catch diodes and back to the DA pins. The fast changing currents and voltage here combined with long PCB traces will cause ringing on the switch pin and may result in unwelcome EMI.
Figure 13. Sample PCB Layout (Top Layer Only)
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TYPICAL APPLICATIONS
1.8V and 3.3V Outputs, Synchronized to 300kHz to 600kHz
VIN = 4.5V TO 36V TRANSIENT TO 60V 2.2F 0.22F SW1 UPS140 CLOCK 12.4k 10k 1.6V 0.4V 22F 22nF DA1 FB1 SYNC 178k
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VIN BOOST1
BD BOOST2 0.22F 15H SW2 LT3509 DA2 FB2 10k 22nF 22F RT GND UPS140 31.6k
VOUT = 1.8V 0.7A
10H
VOUT = 3.3V 0.7A
RUN/SS1 RUN/SS2
NOTE: RT CHOSEN FOR 264kHz
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Automotive Accessory Application 5V Logic Supply and 8V for LCD Display with Display Power Controlled by Logic
TYPICAL APPLICATIONS
VIN = 9.4V TO 36V 2.2F 0.22F SW1 DFLS140L 52.3k DA1 FB1 LT3509 DFLS140L DA2 FB2 10k 90.9k SW2 VIN BOOST1 BD BOOST2 0.22F 10H
VOUT = 5V 0.7A
6.8H
VOUT = 8V 0.7A
RUN/SS1 RUN/SS2 10k 10F 22nF SYNC RT GND 40.2k 10k 10F
0.1F
DISPLAY POWER CONTROL 0V = OFF 3.3V = ON
3509 TA04
fSW = 1MHz
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PACKAGE DESCRIPTION
DE Package 14-Lead Plastic DFN (4mm x 3mm)
(Reference LTC DWG # 05-08-1708 Rev B)
0.70 0.05 3.30 0.05 1.70 0.05 PACKAGE OUTLINE 0.25 0.05 0.50 BSC 3.00 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
3.60 0.05 2.20 0.05
4.00 0.10 (2 SIDES) R = 0.05 TYP
R = 0.115 TYP 8 14
0.40
0.10
3.00 0.10 (2 SIDES) PIN 1 TOP MARK (SEE NOTE 6)
3.30 0.10 1.70 0.10 PIN 1 NOTCH R = 0.20 OR 0.35 45 CHAMFER
(DE14) DFN 0806 REV B
7 0.200 REF 0.75 0.05 3.00 REF 0.00 - 0.05
1 0.25 0.05 0.50 BSC
BOTTOM VIEW--EXPOSED PAD
NOTE: 1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WGED-3) IN JEDEC PACKAGE OUTLINE MO-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
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MSE Package 16-Lead Plastic MSOP Exposed Die Pad ,
(Reference LTC DWG # 05-08-1667 Rev A)
BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 (.112 0.102 .004) 0.889 (.035
PACKAGE DESCRIPTION
0.127 .005)
2.845 (.112 1
0.102 .004) 8 0.35 REF
5.23 (.206) MIN
1.651 (.065
0.102 3.20 - 3.45 .004) (.126 - .136)
1.651 (.065
0.102 .004)
0.12 REF
0.305 0.038 (.0120 .0015) TYP
0.50 (.0197) BSC
16 4.039 0.102 (.159 .004) (NOTE 3)
DETAIL "B" CORNER TAIL IS PART OF DETAIL "B" THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOS
RECOMMENDED SOLDER PAD LAYOUT
16151413121110 9
0.280 0.076 (.011 .003) REF
0.254 (.010) GAUGE PLANE
DETAIL "A" 0 - 6 TYP 4.90 0.152 (.193 .006) 3.00 0.102 (.118 .004) (NOTE 4)
0.53 0.152 (.021 .006) DETAIL "A" 0.18 (.007) 12345678 1.10 (.043) MAX 0.86 (.034) REF
SEATING PLANE
0.50 NOTE: (.0197) 1. DIMENSIONS IN MILLIMETER/(INCH) BSC 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.17 -0.27 (.007 - .011) TYP
0.1016 (.004
0.0508 .002)
MSOP (MSE16) 0608 REV A
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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TYPICAL APPLICATIONS
VIN = 6.5V TO 16V TRANSIENT TO 60V 2.2F 6.8H
2MHz, 5V and 3.3V Outputs
VIN BOOST1 0.1F SW1 MBRM140 52.3k DA1 FB1 LT3509 DA2 FB2 10F 22nF 10k
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BD BOOST2 0.1F SW2 MBRM140 31.6k 4.7H
VOUT = 3.3V 0.7A
VOUT = 5V 0.7A
10F 10k 22nF
RUN/SS1 RUN/SS2 SYNC RT GND 16.9k
fSW = 2MHz
RELATED PARTS
PART NUMBER DESCRIPTION LT1766 LT1936 LT1939 LT1976/ LT1977 LT3434/ LT3435 LT3437 LT3480 LT3481 LT3493 LT3500 LT3501 LT3505 LT3506/ LT3506A LT3507 LT3508 LT3510 LT3684 LT3685 COMMENTS 60V 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT: 1.20V, IQ = 2.5mA, ISD < 25A, TSSOP16/E Package , 36V 1.4A (IOUT) , 500kHz High Efficiency Step-Down DC/DC Converter VIN: 36V to 36V, VOUT: 1.20V, IQ = 1.9mA, ISD < 1A, MS8E Package , 25V 2A, 2.5MHz High Efficiency DC/DC Converter and LDO Controller , 60V, 1.2A (IOUT), 200/500kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode(R) Operation 60V, 2.4A (IOUT), 200/500kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode Operation 60V, 400mA (IOUT),MicroPower Step-Down DC/DC Converter with Burst Mode Operation VIN: 3.6V to 25V, VOUT: 0.8V, IQ = 2.5A, ISD < 10A, 3mm x 3mm DFN-10 VIN: 3.3V to 60V, VOUT: 1.20V, IQ = 100A, ISD < 1A, TSSOP16E Package VIN: 3.3V to 60V, VOUT: 1.20V, IQ = 100A, ISD < 1A, TSSOP16E Package VIN: 3.3V to 60V, VOUT: 1.25V, IQ = 100A, ISD < 1A, 3mm x 3mm DFN-10, TSSOP-16E Package
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High VIN: 3.6V to 38V, VOUT: 0.78V, IQ = 70A, ISD < 1A, 3mm x 3mm Efficiency Step-Down DC/DC Converter with Burst Mode Operation DFN-10, MSOP-10E Package 34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High VIN: 3.6V to 34V, VOUT: 1.26V, IQ = 50A, ISD < 1A, 3mm x 3mm Efficiency Step-Down DC/DC Converter with Burst Mode Operation DFN-10, MSOP-10E Package 36V, 1.4A(IOUT), 750kHz High Efficiency Step-Down DC/DC Converter 36V, 40Vmax, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO Controller 25V, Dual 3A (IOUT), 1.5MHz High Efficiency Step-Down DC/DC Converter 36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz, High Efficiency Step-Down DC/DC Converter 25V, Dual 1.6A (IOUT), 575kHz,/1.1MHz High Efficiency Step-Down DC/DC Converter 36V 2.5MHz, Triple (2.4A + 1.5A + 1.5A (IOUT)) with LDO Controller High Efficiency Step-Down DC/DC Converter 36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz, High Efficiency Step-Down DC/DC Converter 25V, Dual 2A (IOUT), 1.5MHz High Efficiency Step-Down DC/DC Converter 34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High Efficiency Step-Down DC/DC Converter 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter VIN: 36V to 36V, VOUT: 0.8V, IQ = 1.9mA, ISD < 1A, 2mm x 3mm DFN-6 Package VIN: 3.6V to 36V, VOUT: 0.8V, IQ = 2.5mA, ISD < 10A, 3mm x 3mm DFN-10 VIN: 3.3V to 25V, VOUT: 0.8V, IQ = 3.7mA, ISD = 10A, TSSOP-20E Package VIN: 3.6V to 34V, VOUT: 0.78V, IQ = 2mA, ISD < 2A, 3mm x 3mm DFN-8, MSOP-8E Package VIN: 3.6V to 25V, VOUT: 0.8V, IQ = 3.8mA, ISD = 30A, 5mm x 4mm DFN-16 TSSOP-16E Package VIN: 4V to 36V, VOUT: 0.8V, IQ = 7mA, ISD = 1A, 5mm x 7mm QFN-38 VIN: 3.7V to 37V, VOUT: 0.8V, IQ = 4.6mA, ISD = 1A, 4mm x 4mm QFN-24, TSSOP-16E Package VIN: 3.3V to 25V, VOUT: 0.8V, IQ = 3.7mA, ISD = 10A, TSSOP-20E Package VIN: 3.6V to 34V, VOUT: 1.26V, IQ = 850A, ISD < 1A, 3mm x 3mm DFN-10, MSOP-10E Package VIN: 3.6V to 38V, VOUT: 0.78V, IQ = 70A, ISD < 1A, 3mm x 3mm DFN-10, MSOP-10E Package
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Burst Mode is a trademark of Linear Technology Corporation.
24 Linear Technology Corporation
(408) 432-1900
LT 0109 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
FAX: (408) 434-0507 www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2007

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